Evanescent microwave microscopy probe and methodology

ABSTRACT

The present invention generally relates to an evanescent microwave spectroscopy probe and methods for making and using the same. Some embodiments relate to a probe in electrical communication with sapphire tuning capacitors that are arranged in parallel. Some embodiments relate to using capacitors arranged in this manner to achieve higher Q values. Furthermore, probe can be used in microwave microscopy applications, and for imaging samples thereby.

RELATED APPLICATION DATA

This application is a continuation-in-part of, and hereby claims priority to, U.S. patent application Ser. No. 11/255,497 filed on Oct. 20, 2005, pending; which claims priority to U.S. Provisional Patent Application No. 60/620,592 filed on Oct. 20, 2004; all of which are incorporated by reference in their entireties.

BACKGROUND OF THE INVENTION

The present invention relates to probes for imaging evanescent microwave fields. In some embodiments the present invention relates to achieving high Q values by using sapphire capacitors that are arranged in parallel. Additionally, the present invention relates to methods of making and using such probes. Furthermore, some embodiments relate to using such probes for microwave microscopy, and/or for imaging samples thereby.

Prior work in this area used a shunt series combination. Thus, the maximum Q was solely determined by the resistance of the series R-L-C probe equivalent circuit and tuning network. Additionally, prior methods used calculations based on capacitance arising from the gap between a spherical conducting tip and a perfectly conductive sample surface. As a result, such methods do not accurately predict how the probe reacts in the electric field between it and the sample.

In contrast to prevailing methods, the method of the present invention is independent of the electrical properties of the material. Thus, unlike the prior art, the present invention applies equally well to dielectrics, conductors and superconductors. Furthermore, the method of the present invention, as set forth herein, enables the solution of the classical electrodynamic boundary value problem concerning a superconductor modeled as a dielectric having a complex permittivity with a large negative real part, which can be associated with the persistent current. Still further, in some embodiments the resistance is cut by up to about 50% in comparison to prior microwave probes, which results in higher Q values and correspondingly high sensitivity. Thus, the present invention represents a significant advance in the state of the art.

BRIEF SUMMARY OF THE INVENTION

The present invention relates to near field microscopy and, more particularly to an evanescent microwave microscopy probe for use in near field microscopy and methodology for investigating the complex permittivity of a material through evanescent microwave technology. In one embodiment, the probe comprises a low loss, apertured, coaxial resonator that can be tuned over a large bandwidth by a parallel shunt sapphire tuning network. In one embodiment, the transmission line of the probe utilizes high grade paraffin, offering a low loss tangent and a very close dielectric match within the line. A chemically sharpened probe tip extends slightly past the end aperture of the probe and emits a purely evanescent field. The probe is extremely sensitive, achieving Q values in excess of 0.5×10⁶ and a spatial resolution of 1.0×10⁻⁶ meters.

The physical construction of a probe according to the present invention results in a purely evanescent field emanating from its tip. As a result, when a probe of the present invention is used in quantitative microscopy, it is not necessary to use additional hardware and/or a methodology to separate a propagative component from the field. Probes of the present invention also provide an extremely low loss impedance match to standardized equipment. The low loss coaxial resonator of the present invention theoretically has an infinite bandwidth but in practice its bandwidth is governed by its physical length and the source bandwidth. In the present invention the evanescent mode bandwidth is controlled by the aperture diameter, which is quite large compared with state of the art designs.

The probe of the present invention also utilizes a shunt capacitive tuning network characterized by a low equivalent circuit resistance. As a result, the probe of the present invention provides for large resonant frequency selection range and extremely high Q values.

In some embodiments, the present invention relates to an evanescent microwave microscopy probe, comprising: a center conductor having a first end and a second end, wherein the center conductor comprises a waveguide for microwave radiation; a probe tip affixed to the first end of the center conductor, wherein the tip is capable of acquiring a near-field microwave signal from a sample; an outer shield surrounding the center conductor, wherein the center conductor and outer shield are in a generally coaxial relationship, wherein the outer shield has a first end and a second end corresponding to the first and second ends of the center conductor, and wherein the center conductor and outer shield are not in direct contact and thereby form a gap; an insulating material occupying at least a portion of the gap between the center conductor and the outer shield; an aperture located near the tip, wherein the aperture comprises a plate having an inside face and an outside face, wherein the aperture is oriented generally perpendicular to the center conductor, and wherein the aperture comprises a hole that allows the tip to be in microwave communication with a sample; and a tuning network in electronic communication with the second end of the center conductor and with the outer shield, wherein the tuning network comprises a pair of capacitors in a parallel electronic relationship.

The present invention also relates to a process for making a microwave probe comprising: providing a center conductor having a first end and a second end, wherein the center conductor comprises a waveguide for microwave radiation; affixing a probe tip to the first end of the center conductor, wherein the tip is capable of acquiring a near-field microwave signal from a sample; surrounding the center conductor with an outer shield, wherein the center conductor and outer shield are in a generally coaxial relationship, wherein the outer shield has a first end and a second end corresponding to the first and second ends of the center conductor, and wherein the center conductor and outer shield are not in direct contact and thereby form a gap; occupying at least a portion of the gap between the center conductor and the outer shield with an insulating material; providing an aperture located near the tip, wherein the aperture comprises a plate having an inside face and an outside face, wherein the aperture is oriented generally perpendicular to the center conductor, and wherein the aperture comprises a hole that allows the tip to be in microwave communication with a sample; and providing a tuning network in electronic communication with the second end of the center conductor and with the outer shield, wherein the tuning network comprises a pair of capacitors in a parallel electronic relationship.

Further, the present invention relates to a method for detecting a sample using an electromagnetic microwave field comprising: providing the probe of claim 1; obtaining a resonant frequency reference reading from the probe, wherein the probe is substantially decoupled from a sample; placing the probe of claim 1 in electromagnetic microwave communication with the sample; obtaining an resonant frequency reading from the probe; calculating a resonant frequency change relative to the reference reading; and relating the resonant frequency change to one or more properties of the sample.

Still further, the present invention relates to a method for imaging a sample using an electromagnetic microwave field comprising: providing the probe of claim 1; providing an X-Y sample stage, and a sample disposed thereon; obtaining a resonant frequency reference reading from the probe, wherein the probe is substantially decoupled from the sample; placing the probe of claim 1 in electromagnetic microwave communication with the sample at a first position; obtaining a resonant frequency reading from the probe at the first position; moving the probe to a next position; obtaining a resonant frequency reading from the probe at the next position; repeating the preceding two steps as needed; calculating a resonant frequency change at each position relative to the reference reading; and plotting an image of the sample as a function of position and frequency change or a property related to frequency change.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic representation of a cross-section of a probe in accordance with the present invention;

FIG. 2 is a block diagram of a microscope in accordance with the present invention;

FIG. 3 is a diagram of the probe and coupling network;

FIG. 4 is a diagram showing a method of images;

FIG. 5 is a scanning electron micrograph of a superconducting film having two distinct regions;

FIG. 6 is a plot of susceptibility loss versus temperature for a superconducting film;

FIG. 7 is a pair of plots of resonant frequency versus distance between the probe tip and the sample, wherein the data is collected at 79.4 K and 298 K;

FIG. 8 is a plot showing the change in Q for the superconducting film at 79.4 K;

FIG. 9 is a photograph of an embodiment of a microwave microscopy apparatus in accordance with the present invention;

FIG. 10 is a photograph of Ti—Au lines etched on sapphire at 20× magnification;

FIG. 11 is a plot of a change in Q;

FIG. 12 is a plot of a change in reflection coefficient images;

FIG. 13 is a circuit diagram representing a probe connected to a superconductor;

FIG. 14 is a plot showing a change in Q for a superconducting film in junction area of 6° bi-crystal;

FIG. 15 is a plot showing the tuned resonance with a probe tip one micron above a SrTiO₃ crystal sample at 300 K;

FIG. 16 is a plot showing the frequency-shifted resonance with a probe tip about 1 micron from a SrTiO₃ crystal sample at 302 K; and

FIG. 17 is a drawing of a chamfered aperture having a ceramic coating in place of a poly(tetrafluoroethylene) plug.

DETAILED DESCRIPTION OF THE INVENTION

The present invention generally relates to a microwave probe for microwave microscopy and to a method of using the same for generating high quality microwave data. More particularly, the apparatus and method of the present invention can be used to take high-precision, low-noise, measurements of material parameters such as permittivity, permeability, and conductivity.

The probe can be used for the characterization of local electromagnetic properties of materials. The resonator-intrinsic, spatial resolution is experimentally demonstrated herein. A first-order estimation of the sensitivity related to the probe tip-sample interaction for conductors, dielectrics, and superconductors is provided. An estimation of the sensitivity inherent to the resonant probe is set forth. The probe is sensitive in a range of theoretically estimated values, and has micrometer-scale resolution.

In the field of evanescent microwave microscopy, the tip of the probe operates in close proximity to the sample, where the tip radius and effective field distribution range are much smaller than the resonator excitation wavelength. The propagating field exciting resonance in the probe can be ignored and the probe tip-sample interaction can be treated as quasi-static. This can be used for localized measurements and images with resolved features governed essentially by the characteristic size of the tip. The field distribution from the probe tip extends outward a short distance, and as the sample enters the near field of the tip, it interacts with the evanescent field, thus perturbing the probe's resonance. This perturbation is linked to the resonant structure of the probe through the air gap coupling capacitance C_(C) between the tip and the material. This results in the loading of the resonant probe and alters the resonant frequency f_(r), quality factor Q, and reflection coefficient S₁₁ of the resonator.

If the air gap distance from tip to sample is held constant, the f_(r), Q, and S₁₁ variations related to the microwave properties of the sample can be mapped as the probe tip is scanned over the sample. The microwave properties of a material are functions of permittivity ∈, permeability μ, and conductivity σ.

Referring to FIG. 1, the microwave probe 10 of the present invention can be constructed from a 0.085″ semi-ridged coaxial transmission line. The probe 10 is based on an end-wall aperture coaxial transmission line, where the resonator behaves as a series resonant circuit for odd multiples of λ/4.

In constructing probe 10, the center conductor is removed along with the poly(tetrafluoroethylene) insulator and replaced with, for example, high purity paraffin 14. However, the invention is not restricted to paraffin and alternative materials can be used. For example, alternative materials within the scope of the present invention include, without limitation, magnesium oxide, titanium oxide, boron nitride, aluminas. Other waveguide materials include copper, aluminum, brass, alum, and any combination thereof. Still other wave guide materials include polytetrafluoroethylene (PTFE), PTFE/glass fabrics such as Taconic RF-35, and various organic polymeric materials and composites.

Fashioning probe 10 according to the foregoing paragraph results in coaxial wave guide probe 10 rather than an open cavity. A copper aperture, having a thickness of about 0.010″, is soldered inside outer shield 15, creating end-wall aperture 12. Chemically sharpened probe tip 17 is mounted on center conductor 16 and electroplated with silver. The transmission line resonator is then reconstructed by casting the sharpened, plated, center conductor 16 inside the outer shield 15 with high purity paraffin 14. A short section of the original poly(tetrafluoroethylene) shielding replaces paraffin 14 at the sharpened end of the coax, and is located directly above end-wall aperture 12. Poly(tetrafluoroethylene) plug 18 is used to maintain tip-aperture alignment. Sharpened probe tip 17 of center conductor 16 extends beyond shielded end-wall aperture 12 of resonator by approximately 0.001″ or less. The purely evanescent probing field radiates from sharpened probe tip 17. In this manner, as the radius of center conductor 16 decreases, the spatial resolution of the probe increases due to localization of the interaction between probe tip 17 and sample 20.

In an alternative embodiment, poly(tetrafluoroethylene) plug 18, which is disposed at the aperture end, can be replaced with a ceramic, for instance a ceramic coating on the backside of the aperture. Such a coating is applied before aperture 12 is soldered to outer shield 15. In one embodiment this is done using a high temperature strain gauge ceramic adhesive, known by the trade-name Ceramabond-671. In other embodiments the coating can be formed through pulsed laser deposition of any of a wide variety of suitable ceramics including, without limitation, cerium oxide. In one embodiment, the coated side of the aperture is optionally chamfered at about 60° prior to coating (see FIG. 17). Alternatively, the chamfer can be between about 45° and about 75°. This practice results in an increase in Q, and reduced reflection.

Referring to FIG. 2, the microwave excitation frequency of resonant probe 10 can be varied over a bandwidth from about 1 to 40 GHz in network analyzer 40, and is tuned by external capacitors 30. As further illustrated in FIG. 3, microscope probe 10 can be coupled to network analyzer 40 through tuning network capacitors C₁ 31 and C₂ 32, which are connected to center conductor 16 and to outer shield 15.

A block diagram of the microwave microscopy system is shown in FIG. 2. The changes in the probe's resonant frequency, quality factor (Q), and reflection coefficient are tracked by a Hewlett-Packard 8722ES network analyzer 40 through S₁₁ port measurements, as probe 10 moves above sample surface 20. The microwave excitation frequency of resonant probe 10 can be varied within the bandwidth of network analyzer 40 and tuned to critical coupling with tuning assembly 30. Tuning assembly 30 comprises two variable 2.5 to 8 pF capacitors 31, 32. The tuning network has one capacitor C₁ 31 connected in-line with center conductor 16, and the other capacitor C₂ 32 is connected from center conductor 16 to ground.

The X-Y axis stage 70 is driven by Coherent® optical encoded DC linear actuators. Probe 10 is frame-mounted to a Z-axis linear actuator assembly and the height at which probe 10 is above sample surface 20 can be precisely set. The X-Y stage actuators, network analyzer 40, and data acquisition and collection are controlled by computer 50. The program that interfaces to the X-Y stage actuators, serial port communications, 8722ES GPIB interface, and data acquisition is written in National Instruments Labview® software. The complete evanescent microwave scanning system is mounted on a vibration-dampening table (see FIG. 9).

According to one embodiment of the present invention, external tuning capacitor assembly 30 comprises two thermally compensated sapphire capacitors in a shunt configuration. If a shunt is placed near the end of the resonator then the Q of the resonator will theoretically approach infinity. Sapphire capacitors are advantageous because they exhibit frequency invariance up to approximately 10 GHz. In one embodiment, the capacitors 31, 32 are variable from, for example, about 4.5 to 8.0 Picofarads. The position of capacitors 31, 32 in tuning assembly 30 is optimized to reduce interaction. Shielding techniques can also be employed to limit external interaction and leakage.

As is noted above, the present invention also relates to methodology for investigating the complex permittivity of a material through evanescent microwave technology. More particularly, the methodology taught herein is a scheme for investigating the complex permittivity of a material, independent of its other electrical properties, through evanescent microwave spectroscopy.

The extraction of quantitative data through evanescent microwave microscopy requires a detailed configuration of the field outside the probe-tip region. The solution of this field relates the perturbed signal to the distance between the probe-tip and sample, and physical material properties. In accordance with the present invention, the mode of the field generated at the tip is evanescent. A mixed mode field including evanescent and propagative components impedes quantitative measurements. The propagative wave's contribution to the analytical signal depends on the electrical properties of the sample, and limits resolution.

In analyzing conductors quantitatively the probe tip can be modeled as a conducting sphere and the sample as an ideal conductor. The separation between the tip and sample can be modeled as a capacitor with capacitance C_(c), resulting in a resonant frequency shift that is proportional to the variation in C_(c). When a conducting material is placed near the tip electrical interaction therewith causes charge and field redistribution. The method of images can be applied to model this redistribution of the field using a series iteration of two image charges. This variation in tip-sample capacitance results in a detectable shift in resonant frequency of probe 10.

As noted above, the method of images can be applied to quantitatively analyze dielectric materials. Additionally, the probe tip is modeled as a charged conducting sphere with potential V₀. When the tip is placed close to a dielectric material the dielectric is polarized by the tip's electric field. This dielectric reaction to the tip causes a redistribution of charge on the tip in order to maintain the equipotential surface of the sphere, and also results in a shift in the probe's frequency. The method of images can be applied to model the field redistribution using a series of three image charges in an iterative process to meet boundary conditions at the tip of probe 10 and dielectric sample surface 20.

In this unified approach, perturbation theory for microwave resonators is applied to the field distribution outside the tip. The expression for the resonant frequency shift due to the presence of a sample material can be written as $\begin{matrix} {{\frac{\Delta\quad f}{f_{0}} = {{- \frac{\int_{V}{\left\lbrack {{\left( {\Delta\quad ɛ} \right)\left( {\overset{\_}{E} \cdot {\overset{\_}{E}}_{0}} \right)} + {\left( {\Delta\quad\mu} \right)\left( {\overset{\_}{H} \cdot {\overset{\_}{H}}_{0}} \right)}} \right\rbrack{\mathbb{d}V}}}{\int_{V}{\left( {{ɛ_{0}{\overset{\_}{E}}_{0}^{2}} + {\mu_{0}{\overset{\_}{H}}_{0}^{2}}} \right){\mathbb{d}V}}}} = \frac{f - f_{0}}{f_{0}}}},} & (1) \end{matrix}$ where E and H are the perturbed fields, V is the volume of a region outside the resonator tip, f is the resonant frequency and f₀ is the reference frequency. The unperturbed field can be obtained by $\begin{matrix} {{{{E_{0}\left( {r,z} \right)} = {\frac{q}{4{\pi ɛ}_{0}}\frac{\left\lbrack {{r\quad\hat{r}} + {\left( {z + {a_{1}^{\prime}r_{0}}} \right)\hat{z}}} \right\rbrack}{\left\lbrack {r^{2} + \left( {z + {a_{1}^{\prime}r_{0}}} \right)^{2}} \right\rbrack^{3/2}}}},\quad{{{\overset{\_}{H}}_{0}} = {\sqrt{\frac{ɛ_{0}}{\mu_{0}}}{{\overset{\_}{E}}_{0}}}}}{where}} & (2) \\ {{a_{1}^{\prime} = {r_{0} + g}},} & (3) \end{matrix}$ and where r₀ is the radius of the spherical tip and g is the gap between the tip and surface of sample 20. The potential V₀ on the spherical tip is given by $\begin{matrix} {V_{0} = {\frac{q}{4{\pi ɛ}_{0}r_{0}}.}} & (4) \end{matrix}$

By using the method of images (see FIG. 4), the perturbed electric field in the region between the tip and sample, and within the sample volume (assuming r₀ is much smaller than the sample thickness) can be modeled as $\begin{matrix} {{{{\overset{\_}{E}}_{1}\left( {r,z} \right)} = {\frac{q}{4{\pi ɛ}_{0}}{\sum\limits_{n = 1}^{\infty}{q_{n}\left\{ {\frac{\left\lbrack {{r\quad\hat{r}} + {\left( {z + {a_{n}^{\prime}r_{0}}} \right)\hat{z}}} \right\rbrack}{\left\lbrack {r^{2} + \left( {z + {a_{n}^{\prime}r_{0}}} \right)^{2}} \right\rbrack^{3/2}} - {b\frac{\left\lbrack {{r\quad\hat{r}} + {\left( {z - {a_{n}^{\prime}r_{0}}} \right)\hat{z}}} \right\rbrack}{\left\lbrack {r^{2} + \left( {z - {a_{n}^{\prime}r_{0}}} \right)^{2}} \right\rbrack^{3/2}}}} \right\}}}}},{{{\overset{\_}{H}}_{1}} = {\sqrt{\frac{ɛ_{0}}{\mu_{0}}}{{\overset{\_}{E}}_{1}}}},} & (5) \\ {{{{\overset{\_}{E}}_{2}\left( {r,z} \right)} = {\frac{q}{4{\pi\left( {ɛ + ɛ_{0}} \right)}}{\sum\limits_{n = 1}^{\infty}{q_{n}\frac{\left\lbrack {{r\quad\hat{r}} + {\left( {z + {a_{n}^{\prime}r_{0}}} \right)\hat{z}}} \right\rbrack}{\left\lbrack {r^{2} + \left( {z + {a_{n}^{\prime}r_{0}}} \right)^{2}} \right\rbrack^{3/2}}}}}},\quad{{{\overset{\_}{H}}_{2}} = {\sqrt{\frac{ɛ}{\mu}}{{\overset{\_}{E}}_{2}}}},} & (6) \end{matrix}$ where μ is real and $\begin{matrix} {{a_{n}^{\prime} = {a_{1}^{\prime} - \frac{1}{a_{1}^{\prime} + a_{n - 1}^{\prime}}}},\quad{q_{n} = {t_{n}q}},\quad{t_{n} = \frac{{bt}_{n - 1}}{a_{1}^{\prime} + a_{n - 1}^{\prime}}},{t_{1} = 1},\quad{b = \frac{ɛ - ɛ_{0}}{ɛ + ɛ_{0}}},\quad{ɛ = {ɛ^{\prime} + {{\mathbb{i}}\quad{ɛ^{''}.}}}}} & (7) \end{matrix}$

Importantly, for a tip in free space ∈=∈₀ and μ=μ₀ at the location r=0 and z=−g−r₀, E ₀= E ₁= E ₂ and H ₀= H ₁= H ₂, confirming the asymptotic behavior in equations (2), (5), and (6). By integrating the unperturbed electric field in equation (2) and the perturbed electric fields in equations (5) and (6) over a region V outside the spherical tip the frequency shift of equation (1) becomes $\begin{matrix} {{\left( \frac{\Delta\quad f}{f_{0}} \right)_{TOTAL} = {\left( \frac{\Delta\quad f}{f_{0}} \right)_{1} + \left( \frac{\Delta\quad f}{f_{0}} \right)_{2} - {A{\sum\limits_{n = 1}^{\infty}{t_{n}\left\{ {1 - {\frac{1}{2}\left( {1 - b} \right)\frac{1}{a_{1}^{\prime} + a_{n - 1}^{\prime}}}} \right\}}}} - {{A\left( \frac{\Delta\mu}{\Delta ɛ} \right)}\sqrt{\frac{ɛ}{\mu}}\sqrt{\frac{ɛ_{0}}{\mu_{0}}}{\sum\limits_{n = 1}^{\infty}{t_{n}\frac{b}{a_{1}^{\prime} + a_{n - 1}^{\prime}}}}}}},\quad\left( {A = A^{\prime}} \right),{where}} & (8) \\ {{{\left( \frac{\Delta\quad f}{f_{0}} \right)_{1} = {{- A^{\prime}}{\sum\limits_{n = 1}^{\infty}{t_{n}\left\{ {1 - {\frac{1}{2}\left( {1 - b} \right)\frac{1}{a_{1}^{\prime} + a_{n - 1}^{\prime}}}} \right\}}}}},\quad{{Reg}.\quad A},\quad{{\Delta\mu} = 0}}{and}} & (9) \\ {{\left( \frac{\Delta\quad f}{f_{0}} \right)_{2} = {{- {A\left( {1 + {\frac{\Delta\mu}{\Delta ɛ}\sqrt{\frac{ɛ}{\mu}}\sqrt{\frac{ɛ_{0}}{\mu_{0}}}}} \right)}}{\sum\limits_{n = 1}^{\infty}{t_{n}\frac{b}{a_{1}^{\prime} + a_{n - 1}^{\prime}}}}}},\quad{{Reg}.\quad B.}} & (10) \end{matrix}$

Parameters A and A′ are constants determined by the geometry of the tip-resonator assembly. Taking into account the real part of equation (8), the analytical expression, fits with the experimental data below.

As noted above, prior methods used calculations based on capacitance arising from the gap between a spherical conducting tip and a perfectly conductive sample surface. As a result, such methods do not accurately predict how the probe reacts in the electric field between it and the sample. The method of the present invention overcomes this deficiency. Moreover, the results of the prior art can be reproduced by the present method if the following additional restrictions are imposed on the reaction of the resonator probe to electric fields outside the tip. Namely, the coefficients in equations (9) and (10) must be equal (A′=A). This assumption provides a smooth transition between insulators and ideal conductors by assuming b=1 in equation (8).

In one embodiment the method of the present invention is used to measure the dielectric properties of the superconductor YBa₂Cu₃O_(7-δ). A superconductor can be treated as a dielectric material with a negative dielectric constant rather than a low loss conductor. In this embodiment probe 10 comprises a tuned, end-wall apertured coaxial transmission line. Resonator probe 10 is coupled to network analyzer 40 through tuning network 30 and coupled to sample 20 (see FIG. 2). When probe tip 17 is in close proximity to sample 20, the resonator's frequency f shifts. In measuring the frequency shift, the resonant frequency reference is set with probe tip 17 at a fixed distance above sample 20. The distance between probe tip 17 and sample 20 is sufficient to assure that the evanescent field emanating from tip 17 will not interact with sample 20. The field dispersion from the probe tip extends outward a short distance with the amplitude of the evanescent field decaying exponentially. As sample 20 enters the near field of probe tip 17 it interacts with the evanescent field, thereby perturbing it. This results in loading the probe 10. Accordingly, sample 20 is considered part of the resonant circuit and results in losses to the system, which decreases the probe's 10 resonant frequency. The measured frequency shift, as it relates to tip-sample separation g, generates a transfer function relating Δf to Δg. The transfer function is best fit with an electrostatic field model using the method of images to extract complex permittivity values.

In one variation of the foregoing embodiment, the evanescent microwave microscopy system is adapted for making cryogenic measurements. A miniature single-stage Joule-Thompson cryogenic system is fixed to X-Y stage 70. Microwave probe 10 is fitted through a bellows, which provides a vacuum seal and allows the probe to move freely over sample 20, which is mounted on a cryogenic finger directly below probe 10.

In this embodiment, an YBa₂Cu₃O_(7-δ) superconducting thin film is fabricated by pulsed laser deposition. This deposition method results in two distinct regions, 1 and 2, forming on a 0.5 mm thick LaAlO₃ substrate (see FIG. 5). The superconductive transition temperatures for region 1 and 2 of the film are T_(c)=92 K and 90 K respectively, which are measured by plotting susceptibility loss versus temperature under different amplitudes of alternating magnetic field at the frequency of 2 MHz, as shown in FIG. 6. The measured frequency shift data is collected for both regions at 79.4 K and 298 K as shown in FIG. 7. Fitting parameters from equation (8) to the experimental data are consolidated in Table I. TABLE I SIMULATION FIT PARAMETERS FOR YBa₂Cu₃O₇ _(τ) _(δ) SUPERCONDUCTING THIN FILM AT 79.4 K AND 298 K. A ε′/ε₀ r₀ μ/μ₀ REGIONS (10⁻⁴) (10⁸) ε″/ε₀ (10⁻⁶m) (10⁻⁴) REGION 1 at 79.4 K 2.09 −9.2 −0.1 3.35 1 REGION 2 at 79.4 K 2.08 −9 −0.1 3.35 1 TRANSITION REGION 2.08 −9.1 −0.1 3.35 1 at 79.4 K REGION 1 at 298 K 1.45 1 6.6 8 1 REGION 2 at 298 K 1.45 1 6.85 8 1

Above the transition temperature (T_(c)), the superconductor behaves like a metallic conductor. Thus, the sign and magnitude of the real and imaginary permittivity values change (Table I). FIG. 7 shows the curves from both regions below T_(c) and illustrates that there is a distinct measurable difference between these regions. The transition section connecting regions 1 and 2 with the associated frequency shift fit parameters generated at 79.4 K falls in between fit curves for regions 1 and 2. The model fit parameters for this transition segment are A=2.08×10⁻⁴, which is the resonator scaling factor, the real component of permittivity ∈=−9.13×10⁸ _(∈) ₀ , the imaginary component of permittivity ∈″=−0.1_(∈) ₀ , and the effective tip radius r₀=3.35 μm. FIG. 8 shows a change in Q scan performed at 79.4 K over both regions, and indicates the average dynamic range of Q in this scan between the two areas is approximately 8000. The higher Q level is associated with the area of T_(c)=92 K, and the lower Q level corresponds to region of T_(c)=90 K.

The resolution of the probe is verified using a sapphire polycrystalline substrate with titanium-gold etched lines of widths ranging from about 10 μm to 1 μm (see FIG. 10). Titanium is used for adhesion of the gold to the substrate, and is approximately 100 nm thick, while the thickness of the gold deposition is approximately 1 μm. The resonant frequency of the probe is tuned to 2.67 GHz. The etched lines of the sample are scanned with the probe resulting in a change in frequency, Q, and magnitude of reflection plots.

The smallest physically resolvable feature for an evanescent probe is governed by the size of the tip radius, along with the height at which the tip is positioned above the feature. For example, to resolve a 5 μm physical feature, the probe tip radius r₀ must be approximately 5 μm or less from tip to sample.

The change in Q and change in magnitude of reflection coefficient images are illustrated in FIGS. 11 and 12, respectively. The data for these plots are taken from a 20 μm×18 μm scan area around a 1 μm wide etched line. The measured tip radius of the probe used is 1.2 μm with a stand off height (g) of 2 μm and a 1 μm data acquisition step. The location of the etched line is indicated on each plot by arrows with corresponding measurements in micrometers. In this embodiment, the one micrometer line was distinguishable in both plots, which gives the probe about 1 μm topographical resolution or better. The Q values attainable with this tunable probe range from 1.5×10⁴ to well over 10⁵, and even over 10⁶. According to this embodiment, the dynamic range of the change in Q is approximately 5×10⁵ , as shown in FIG. 11.

The Johnson noise-limited sensitivity is analyzed in the present invention by setting the signal power equal to the noise power resulting in [(δ∈/∈)]=2.45×10⁻⁵. As those of ordinary skill in the art are aware, Johnson noise results from random thermal movements of charge carriers, and is often referred to alternatively as thermal noise.

The sensitivity of the evanescent microwave probe described herein can be separated into two categories. The first, S_(r), is inherent to the resonator itself and directly proportional to it's quiescent operating value Q. The other, S_(f), is external to the resonator and solely determined by tip-sample interactions. A noise threshold has to be considered in an evanescent microwave system, which also affects sensitivity.

The minimum detectable signal in an evanescent microwave microscopy system should be greater than the noise created by the probe, tuning network, and coupling to the sample. The noise is generated by a resistance at an absolute temperature of T by the random motion of electrons proportional to the temperature T within the resistor. This generates random voltage fluctuations at the resistor terminal, which has a zero average value, but a non-zero rms value given by Planck's'black body radiation law. These voltage fluctuations can be calculated by the Raleigh-Jeans approximation as V _(n(rms))=√{square root over (4kTBR)},  (11) where k=1.38×10⁻²³ J/K is Boltzmann's constant, T is the temperature in Kelvin, B is the bandwidth of the system in Hertz, and R is the resistance in ohms. The resistance that results at critical coupling is the resistance R that produces noise in the system. Therefore, the signal level should be above this noise level in order to be detectable.

The sensitivity approximation internal to the resonator S_(r) can be determined theoretically and experimentally. The theoretical value is analytically approximated by considering the lumped series equivalent circuit of the resonator, which has an inherent resonant frequency ω₀ and Q associated with the lumped parameters R₀, L₀, and C₀. This configuration and associated parameters can be viewed as if the probe tip is beyond the decay length of the evanescent field from a material, or in free space. If the probe tip is brought into close proximity and electrically couples to the sample, the resonant frequency ω₀ and Q are perturbed to a new value ω′₀ and Q′, respectively, and are associated with new perturbed parameters R′₀, L′₀, and C′₀. The total impedance looking into the terminals of the perturbed resonator coupled to a sample can be written as $\begin{matrix} {Z_{TOTAL} = {{R_{0}^{\prime}\left\lbrack {1 + {j\quad{Q\left( {\frac{\omega}{\omega_{0}^{\prime}} - \frac{\omega_{0}^{\prime}}{\omega}} \right)}}} \right\rbrack}.}} & (12) \end{matrix}$

The magnitude of the reflection coefficient S₁₁ is related to Z_(TOTAL) by $\begin{matrix} {{S_{11} = \frac{Z_{TOTAL} - Z_{0}}{Z_{TOTAL} + Z_{0}}},} & (13) \end{matrix}$ where Z₀ is the characteristic impedance of the resonant structure. If one assumes critical coupling, where the resonator is matched to the characteristic impedance of the feed transmission line at resonant frequency, then R′₀≈Z₀ at ω≈ω′₀ and S_(r) is defined in as $\begin{matrix} {{S_{r} = {\frac{\mathbb{d}S_{11}}{\mathbb{d}\omega} \approx {\frac{Q^{\prime}}{\omega_{0}^{\prime}}\left( {1 - \frac{\Delta\omega}{\omega_{0}^{\prime}}} \right)}}},} & (14) \end{matrix}$ where Δω=ω−ω′₀.

The external sensitivity determined by tip-sample interaction of probe 10 is based on a λ/4 section of transmission line, with the lumped parameter series equivalent circuit coupled to an equivalent circuit model of a superconductor shown in FIG. 7. The series lumped parameter circuit for the resonator consists of R₀, L₀, and C₀ and the probe tip coupling to the superconductor is represented by C_(C). The equivalent circuit model of the superconductor is comprised of R_(S), L_(S), C_(S), and L_(C), where the series combination of R_(S) and L_(S) represents the normal conduction. The element L_(C) signifies the kinetic inductance of the Cooper-pair flow and C_(S) is related to displacement current. The superconductor equivalent circuit contains the necessary circuit elements in the appropriate configuration to represent not only a superconductor, but a metallic conductor and a dielectric.

The equivalent circuit model for the probe coupled to a superconductor is illustrated in FIG. 7, where the equivalent circuit model for the superconductor is derived from the two-fluid model. The lumped circuit representation of the superconductor comprises capacitance C_(S), the inductance for normal carrier flow L_(S), and resistivity ρ=1/σ₁, shunted by kinetic inductance L_(C)=1/ωσ₂. The parameters C_(S) and L_(S) are considered to have minimal effects when the superconductor is subjected to low frequencies and is neglected in this analysis. The conductivity ratio y=σ₁/σ₂ is correlated to the impedance ratio y=ωL_(C)/ρ and in the limit of large y (y>>1), σ₂=0 and L_(C)>>1. The opposite extreme, y<<1 results in L_(C) approaching 0, while σ₂ advances toward infinity. The superconductive samples for this study were subjected to a frequency of approximately 1 GHz and are of an inductive nature. The superconductor with an inductive nature has L_(C)<<R_(S).

The impedance Z₁ is the parallel combination of R_(S) and L_(C) and is represented as $\begin{matrix} {Z_{1} = {\frac{{j\omega}\quad L_{C}R_{S}}{R_{S} + {{j\omega}\quad L_{C}}}.}} & (15) \end{matrix}$

The impedance Z₂ is the series combination of C_(C) and Z_(1,) which results in $\begin{matrix} {Z_{2} = {{\frac{1}{{j\omega}\quad C_{C}} + \frac{{j\omega}\quad L_{C}R_{S}}{R_{S} + {{j\omega}\quad L_{C}}}} = {\frac{R_{S} + {{j\omega}\quad L_{C}} + {{j\omega}\quad{C_{C}\left( {{j\omega}\quad L_{C}R_{S}} \right)}}}{{j\omega}\quad{C_{C}\left( {R_{S} + {{j\omega}\quad L_{C}}} \right)}}.}}} & (16) \end{matrix}$

The impedance Z₃ is the parallel combination of Z₂ and C₀ given by $\begin{matrix} {\begin{matrix} {\frac{1}{\quad Z_{\quad 3}} = {\frac{1}{\quad Z_{\quad 2}} + {{j\omega}\quad C_{\quad 0}}}} \\ {= {\frac{{j\omega}\quad{C_{C}\left( {R_{S} + {{j\omega}\quad L_{C}}} \right)}}{R_{S} + {{j\omega}\quad L_{C}} + {{j\omega}\quad{C_{C}\left( {{j\omega}\quad L_{C}R_{S}} \right)}}} + {{j\omega}\quad C_{0}}}} \end{matrix}\begin{matrix} {Z_{3} = \frac{R_{S} - {\omega^{2}L_{C}C_{C}R_{S}} + {{j\omega}\quad L_{C}}}{{{j\omega}\quad{C_{C}\left( {R_{S} + {{j\omega}\quad L_{C}}} \right)}} + {C_{0}\left( {R_{S} - {\omega^{2}L_{C}C_{C}R_{S}} + {{j\omega}\quad L_{C}}} \right)}}} \\ {= {- {\frac{j}{\omega}\left\lbrack \frac{R_{S} - {\omega^{2}L_{C}C_{C}R_{S}} + {{j\omega}\quad L_{C}}}{{{j\omega}\quad{C_{C}\left( {R_{S} + {{j\omega}\quad L_{C}}} \right)}} + {C_{0}\left( {R_{S} - {\omega^{2}L_{C}C_{C}R_{S}} + {{j\omega}\quad L_{C}}} \right)}} \right\rbrack}}} \\ {= {{- \frac{j}{\omega}}{Z_{3}^{\prime}.}}} \end{matrix}} & (17) \end{matrix}$

The total impedance Z_(TOTAL) looking into the terminals of the probe coupled to a superconductor sample is $Z_{total} = {R_{0} + {{j\omega}\quad L_{0}} - {\frac{j}{\omega}{Z_{3}^{\prime}.}}}$

The complex impedance Z₃ can be represented as $Z_{3} = {{\frac{1}{j\omega}\left\lbrack {{Re}\left( Z_{3}^{\prime} \right)} \right\rbrack} = {- {{\frac{j}{\omega}\left\lbrack {{Re}\left( Z_{3}^{\prime} \right)} \right\rbrack}.}}}$

At resonance, the inductive and capacitive reactances cancel; therefore, $\begin{matrix} {{{{{j\omega}\quad L_{0}} - {\frac{j}{\omega}\left\lbrack {{Re}\left( Z_{3}^{\prime} \right)} \right\rbrack}} = 0},\quad{{\omega^{2}L_{0}} = {{{Re}\left( Z_{3}^{\prime} \right)}.}}} & (18) \end{matrix}$

This allows one to solve for perturbed frequency ω in terms of the perturbed lumped circuit parameters in an iterative process, where one will be taking a first-order approximation. The combination of equations (7) and (8) results in $\begin{matrix} \begin{matrix} {{\omega^{2}L_{0}} = \frac{{R_{S}^{2}\left( {1 - {\omega^{2}L_{C}C_{C}}} \right)}\left( {C_{C} + C_{0} - {\omega^{2}C_{0}L_{C}C_{C}}} \right)}{\left( {C_{C} - C_{0} - {\omega^{2}C_{0}L_{C}C_{C}}} \right)^{2} + {\omega^{2}{L_{C}^{2}\left( {C_{C} + C_{0}} \right)}^{2}}}} \\ {= {\frac{1}{\left( {C_{C} + C_{0}} \right)}\frac{\left( {1 - {\omega^{2}L_{C}C_{C}} - {\omega^{2}L_{C}\frac{C_{0}C_{C}}{C_{C} + C_{0}}}} \right)}{\left( {1 - {2\omega^{2}L_{C}\frac{C_{0}C_{C}}{C_{C} + C_{0}}}} \right)}}} \\ {= {\frac{1}{\left( {C_{C} + C_{0}} \right)}\frac{1}{\begin{matrix} \left( {1 - {2\omega^{\quad 2}L_{\quad C}\frac{\quad{C_{\quad 0}\quad C_{\quad C}}}{\quad{C_{\quad C}\quad + \quad C_{\quad 0}}}}} \right) \\ \left\lbrack {1 + {\omega^{2}L_{C}{C_{C}\left( {1 + \frac{C_{0}}{C_{C} + C_{0}}} \right)}}} \right\rbrack \end{matrix}}}} \\ {= {\frac{1}{\left( {C_{C} + C_{0}} \right)}{\frac{1}{1 + {\omega^{2}L_{C}\frac{C_{C}^{2}}{C_{C} + C_{0}}}}.}}} \end{matrix} & (19) \end{matrix}$

Therefore, for the first iteration, one has the following equation $\begin{matrix} {\omega_{0}^{\prime 2} = {\frac{1}{L_{0}\left( {C_{C} + C_{0}} \right)}{\frac{1}{\left\lbrack {1 + {\frac{L_{C}}{L_{0}}\left( \frac{C_{C}}{C_{C} + C_{0}} \right)^{2}}} \right\rbrack}.}}} & (20) \end{matrix}$

Solving for ω′₀ in equation (20) results in $\begin{matrix} {{\omega_{0}^{\prime} = {\omega_{0}\frac{1}{\sqrt{1 + \frac{C_{C}}{C_{0}}}}\frac{1}{\sqrt{1 + \frac{L_{C}}{L_{0}}}\left( \frac{C_{C}}{C_{C} + C_{0}} \right)^{2}}}},} & (21) \end{matrix}$ where $\frac{L_{C}}{L_{0}} ⪡ 1.$

The Taylor expansion of equation (21) gives $\begin{matrix} {\omega_{0}^{\prime} = {{{\omega_{0}\left( {1 - {\frac{1}{2}\frac{C_{C}}{C_{0}}}} \right)}\left\lbrack {1 - {\frac{1}{2}\frac{L_{C}}{L_{0}}\frac{C_{C}^{2}}{\left( {C_{0} + C_{C}} \right)^{2}}}} \right\rbrack}.}} & (22) \end{matrix}$

The sensitivity S_(f) for a superconductor is defined as $\begin{matrix} {{S_{f} = {\frac{g_{S}R_{S}^{2}}{2\pi}{\frac{\mathbb{d}\omega_{0}^{\prime}}{\mathbb{d}L_{C}}}}},} & (23) \end{matrix}$ where ${g_{S} = \frac{A_{eff}}{\lambda\quad L}},$ A_(eff) is the effective tip area, and λ_(L) is the London penetration depth. Therefore, the sensitivity S_(f) for a superconductor is found by taking the derivative of ω′₀ with respect to L_(C) in equation (22) and is given by $\begin{matrix} {S_{f} = {\frac{g_{S}R_{S}^{2}}{2\pi}{{{\omega_{0}\left( {1 - \frac{C_{C}}{2C_{0}}} \right)}\left\lbrack {\frac{1}{\left( {2L_{0}} \right)}\frac{C_{C}^{2}}{\left( {C_{C} + C_{0}} \right)^{2}}} \right\rbrack}.}}} & (24) \end{matrix}$

The ability of the probe to differentiate between regions of different conductivity within a superconductor Δσ/σ is defined as $\begin{matrix} {\frac{\Delta\sigma}{\sigma} = {{\left( \frac{V_{n{({rms})}}}{V_{i\quad n}} \right)/S_{f}}S_{r}{\sigma.}}} & (25) \end{matrix}$

The probe couples to a metallic sample through the coupling capacitance C_(C) and the conductor is represented as the series combination of R_(S) and L_(S). An equivalent circuit of a metallic sample does not contain the circuit elements L_(C) and C_(S) in the two-fluid equivalent circuit (see FIG. 13). Therefore, C_(S)=0 and L_(C)=∞. The impedance Z₁ is the series combination of C_(C), R_(S), and L_(S) and is represented as $\begin{matrix} {Z_{1} = {{R_{S} + {{j\omega}\quad L_{S}} + \frac{1}{{j\omega}\quad C_{C}}} = {\frac{1 + {{j\omega}\quad C_{C}R_{S}} - {\omega^{2}L_{S}C_{C}}}{{j\omega}\quad C_{C}}.}}} & (26) \end{matrix}$

The parallel combination of Z₁ and C₀ results in $\begin{matrix} {\frac{1}{Z_{2}} = {{{j\omega}\quad C_{0}} + \frac{{j\omega}\quad C_{C}}{\left( {1 - {\omega^{2}L_{S}C_{C}}} \right) + {{j\omega}\quad C_{C}R_{S}}}}} \\ {= \frac{{{j\omega}\quad{C_{0}\left( {1 - {\omega^{2}L_{S}C_{C}}} \right)}} + {{j\omega}\quad C_{C}} - {\omega^{2}C_{0}C_{C}R_{S}}}{\left( {1 - {\omega^{2}L_{S}C_{C}}} \right) + {{j\omega}\quad C_{C}R_{S}}}} \\ {{= \frac{{j\omega}\quad\left\lbrack {C_{C} + {C_{0}\left( {1 - {\omega^{2}L_{S}C_{C}}} \right)} - {\omega^{2}C_{0}C_{C}R_{S}}} \right\rbrack}{1 - {\omega^{2}L_{S}C_{C}} + {{j\omega}\quad C_{C}R_{S}}}},} \end{matrix}$ and the impedance Z₂ is $\begin{matrix} {Z_{2} = {\frac{\left( {1 - {\omega^{2}L_{S}C_{C}}} \right) + {{j\omega}\quad C_{C}R_{S}}}{{j\omega}\left\lbrack {{C_{C}{C_{0}\left( {1 - {\omega^{2}L_{S}C_{C}}} \right)}} - {\omega^{2}C_{0}C_{C}R_{S}}} \right\rbrack} = {{- \frac{j}{\omega}}{Z_{2}^{\prime}.}}}} & (27) \end{matrix}$

The total impedance Z_(TOTAL) looking into the terminals of the probe coupled to a conductor sample is $Z_{TOTAL} = {R_{0} + {{j\omega}\quad L_{0}} - {\frac{j}{\omega}{Z_{2}^{\prime}.}}}$

The complex impedance Z₃ can be represented as $Z_{2} = {{\frac{1}{j\omega}\left\lbrack {{Re}\left( Z_{2}^{\prime} \right)} \right\rbrack} = {- {{\frac{j}{\omega}\left\lbrack {{Re}\left( Z_{2}^{\prime} \right)} \right\rbrack}.}}}$

At resonance, the inductive and capacitive reactance cancel; therefore, $\begin{matrix} {{{{{j\omega}\quad L_{0}} - {\frac{j}{\omega}\left\lbrack {{Re}\left( Z_{2}^{\prime} \right)} \right\rbrack}} = 0},\quad{{\omega^{2}L_{0}} = {{Re}{\left( Z_{2}^{\prime} \right).}}}} & (28) \end{matrix}$

The impedance z′₂ is represented as $\begin{matrix} {Z_{2}^{\prime} = {\frac{\left( {1 - {\omega^{2}L_{S}C_{C}}} \right) + {{j\omega}\quad C_{C}R_{S}}}{{j\omega}\left\lbrack {C_{C} + {C_{0}\left( {1 - {\omega^{2}L_{S}C_{C}}} \right)} - {\omega^{2}C_{0}C_{C}R_{S}}} \right\rbrack}.}} & (29) \end{matrix}$

Taking the real part of equation (29), we have $\begin{matrix} \begin{matrix} {{{Re}\left( Z_{2}^{\prime} \right)} = \frac{{\left( {1 - {\omega^{2}L_{S}C_{C}}} \right)\left\lbrack {C_{C} + {C_{0}\left( {1 - {\omega^{2}L_{S}C_{C}}} \right)}} \right\rbrack} + {\omega^{2}C_{0}C_{C}^{2}R_{S}^{2}}}{\left\lbrack {C_{C} + {C_{0}\left( {1 - {\omega^{2}L_{S}C_{C}}} \right)}} \right\rbrack^{2} + {\omega^{2}C_{0}C_{C}^{2}R_{S}^{2}}}} \\ {= {\frac{{C_{C}\left( {1 - {\omega^{2}L_{S}C_{C}}} \right)} + {C_{0}\left( {1 - {\omega^{2}L_{S}C_{C}}} \right)}^{2} + {\omega^{2}C_{0}C_{C}^{2}R_{S}^{2}}}{\left\lbrack {C_{C} + {C_{0}\left( {1 - {\omega^{2}L_{S}C_{C}}} \right)}} \right\rbrack^{2} + {\omega^{2}C_{0}C_{C}^{2}R_{S}^{2}}}.}} \end{matrix} & (30) \end{matrix}$

The numerator and denominator of equation (30) are considered separately, so the numerator is expanded and results in (C_(C)+C₀)−ω²(L_(S)C_(C) ²+2C₀L_(S)C_(C)−C₀C_(C) ²R_(S) ²)+ω⁴C₀C_(C) ²L_(S) ².  (31)

The ω⁴ term in equation (31) is discarded due to insignificance and the denominator of equation (30) is expanded as (C _(C) +C ₀−ω² L _(S) C _(C) C ₀)²+ω² C ₀ ² C _(C) ² R _(S) ²⁼⁽ C _(C) +C ₀)²−2ω² L _(S)(C _(C) +C ₀)C _(C) C ₀+ω⁴ C ₀ ² L _(S) ²+ω² C ₀ ² C _(C) ² R _(S) ².  (32)

Likewise, the ω⁴ term in equation (32) is neglected and the combination of equations (31) and (32) appear as $\begin{matrix} {\frac{\left( {C_{C} + C_{0}} \right) - {\omega^{2}\left( {{L_{S}C_{C}^{2}} + {2C_{0}L_{S}C_{C}} - {C_{0}C_{C}^{2}R_{S}^{2}}} \right)}}{\left( {C_{C} + C_{0}} \right)^{2} - {2\omega^{2}{L_{S}\left( {C_{C} + {C_{0}L_{S}}} \right)}C_{0}C_{C}} + {\omega^{2}C_{0}^{2}C_{C}^{2}R_{S}^{2}}}.} & (33) \end{matrix}$

Factoring out (C_(C)+C₀) in numerator and denominator of equation (33) and substituting the result into equation (28) produces $\begin{matrix} {{\omega^{2}L_{0}} = {\frac{1}{\left( {C_{C} + C_{0}} \right)}{\frac{1 - {\omega^{2}\frac{\left( {{L_{S}C_{C}^{2}} + {2C_{0}L_{S}C_{C}^{2}} - {C_{0}C_{C}^{2}R_{S}^{2}}} \right)}{\left( {C_{C} + C_{0}} \right)}}}{1 - {2\omega^{2}\frac{L_{S}C_{C}C_{0}}{\left( {C_{C} + C_{0}} \right)}} + {\omega^{2}\frac{C_{0}^{2}C_{C}^{2}R_{S}^{2}}{\left( {C_{C} + C_{0}} \right)^{2}}}}.}}} & (34) \end{matrix}$

Reducing equation (34) and multiplying by ${1 + {\omega^{2}\frac{\left\lbrack {{L_{S}{C_{C}\left( {C_{C} + {2C_{0}}} \right)}} - {C_{0}C_{C}^{2}R_{S}^{2}}} \right\rbrack}{\left( {C_{C} + C_{0}} \right)}}},$ results in $\begin{matrix} {{\omega^{2}L_{0}} = {\frac{1}{\left( {C_{C} + C_{0}} \right)}{\frac{1}{1 + {\omega^{2}\frac{L_{S}C_{C}^{2}}{\left( {C_{C} + C_{0}} \right)}} + {\omega^{2}\frac{C_{0}C_{C}^{2}R_{S}^{2}}{\left( {C_{C} + C_{0}} \right)}\left( {\frac{C_{0}}{C_{C} + C_{0}} - 1} \right)}}.}}} & (35) \end{matrix}$

The relation ω₀ ²/(1+C_(C)/C₀) with ω₀ ²=1/L₀C₀ as a zero-order approximation to our iterative process is substituted into equation (35) producing a first-order approximation $\begin{matrix} {\omega_{0}^{\prime 2} = {\frac{1}{L_{0}\left( {C_{C} + C_{0}} \right)}{\frac{1}{1 + {\frac{L_{S}}{L_{0}}\left( \frac{C_{C}}{C_{C} + C_{0}} \right)^{2}} - {\frac{C_{0}R_{S}^{2}}{L_{0}}\left( \frac{C_{C}}{C_{C} + C_{0}} \right)^{3}}}.}}} & (36) \end{matrix}$

Rewriting equation (36) and taking the square root of both sides and neglecting higher-order terms, we have the first-order approximation for the perturbed resonant frequency due to the coupling of the probe to a conductor. $\begin{matrix} {\omega_{0}^{\prime} = {\omega_{0}\frac{1}{\sqrt{1 + \frac{C_{C}}{C_{0}}}}{\frac{1}{\sqrt{1 + {\frac{L_{S}}{L_{0}}\left( \frac{C_{C}}{C_{C} + C_{0}} \right)^{2}}}}.}}} & (37) \end{matrix}$

The Taylor expansion of equation (37) gives $\begin{matrix} {\omega_{0}^{\prime} = {{{\omega_{0}\left( {1 - \frac{C_{C}}{2C_{0}}} \right)}\left\lbrack {1 - {\frac{L_{S}}{2L_{0}}\frac{C_{C}^{2}}{\left( {C_{C} + C_{0}} \right)^{2}}}} \right\rbrack}.}} & (38) \end{matrix}$

The sensitivity S_(f) for a conductor is defined as $\begin{matrix} {{S_{f} = {\frac{g_{S}R_{S}^{2}}{2\pi}{\frac{\mathbb{d}\omega_{0}^{\prime}}{\mathbb{d}L_{S}}}}},} & (39) \end{matrix}$ where ${g_{S} = \frac{A_{eff}}{\delta}},$ A_(eff) is the effective tip area, and δ is the skin depth. Therefore, the sensitivity S_(f) equation (39) for a conductor is found by taking the derivative of ω′₀ with respect to L_(S) in equation (38) and results in $\begin{matrix} {S_{f} = {\frac{g_{S}R_{S}^{2}}{2\pi}{{{\omega_{0}\left( {1 - \frac{C_{C}}{2C_{0}}} \right)}\left\lbrack {\frac{1}{2L_{0}}\frac{C_{C}^{2}}{\left( {C_{C} + C_{0}} \right)^{2}}} \right\rbrack}.}}} & (40) \end{matrix}$

The ability of the probe to differentiate between regions of different conductivity Δσ/σ is defined as $\begin{matrix} {{\frac{\Delta\sigma}{\sigma} = {{\left( \frac{V_{n{({rms})}}}{V_{i\quad n}} \right)/S_{f}}S_{r}\sigma}},} & (31) \end{matrix}$ where v_(n(rms)) is given in equation (11) and v_(in) is the probe input voltage.

The probe also couples to a dielectric sample through the coupling capacitance C_(C) and the dielectric is represented as the parallel combination of R_(S) and C_(S). The equivalent circuit of an insulating sample does not contain the circuit elements L_(C) and L_(S) from the two-fluid equivalent circuit. Therefore, L_(S)=0 and L_(C)=∞. The impedance Z₁ is the parallel combination of R_(S) and C_(S) and is represented as $\begin{matrix} {Z_{1} = {\frac{R_{S}}{{{j\omega}\quad C_{S}R_{S}} + 1}.}} & (42) \end{matrix}$

The series combination of Z₁ and C_(C) result in $\begin{matrix} {Z_{2} = {{\frac{1}{{j\omega}\quad C_{C}} + \frac{1}{{{j\omega}\quad C_{S}R_{S}} + 1}} = \frac{1 + {{j\omega}\quad C_{S}R_{S}} + {{j\omega}\quad C_{C}R_{S}}}{{j\omega}\quad{C_{C}\left( {{{j\omega}\quad C_{S}R_{S}} + 1} \right)}}}} & (43) \end{matrix}$

The impedance Z₃ is the parallel combination of Z₂ and C₀ and is represented as $\begin{matrix} {{\frac{1}{Z_{3}} = {\frac{{j\omega}\quad{C_{C}\left( {1 + {{j\omega}\quad C_{S}R_{S}}} \right)}}{\quad\left( {1 + {{j\omega}\quad C_{C}R_{S}} + {{j\omega}\quad C_{S}R_{S}}} \right)} + {{j\omega}\quad C_{0}}}},\begin{matrix} {Z_{\quad 3} = \frac{1 + {{j\omega}\quad C_{C}R_{S}} + {{j\omega}\quad C_{S}R_{S}}}{{{j\omega}\quad{C_{C}\left( {1 + {{j\omega}\quad C_{S}R_{S}}} \right)}} + {{j\omega}\quad{C_{0}\left( {1 + {{j\omega}\quad C_{C}R_{S}} + {{j\omega}\quad C_{S}R_{S}}} \right)}}}} \\ {= {{- \frac{j}{\quad\omega}}{Z_{\quad 3}^{\quad\prime}.}}} \end{matrix}} & (44) \end{matrix}$

The total impedance Z_(TOTAL) looking into the terminals of the probe coupled to a dielectric sample is $Z_{TOTAL} = {R_{0} + {{j\omega}\quad L_{0}} - {\frac{j}{\omega}{Z_{3}^{\prime}.}}}$

The complex impedance Z₃ can be represented as $Z_{3} = {{\frac{1}{j\omega}\left\lbrack {{Re}\left( Z_{3}^{\prime} \right)} \right\rbrack} = {- {{\frac{j}{\omega}\left\lbrack {{Re}\left( Z_{3}^{\prime} \right)} \right\rbrack}.}}}$

At resonance, the inductive and capacitive reactance cancel; hence, $\begin{matrix} {{{{{j\omega}\quad L_{0}} - {\frac{j}{\omega}\left\lbrack {{Re}\left( Z_{3}^{\prime} \right)} \right\rbrack}} = 0},\quad{{\omega^{2}\quad L_{0}} = {{Re}{\left( Z_{3}^{\prime} \right).}}}} & (45) \end{matrix}$

The quantity jωR_(S) is factored out in the numerator and denominator of equation (44) and the result is placed into equation (45), giving $\begin{matrix} {{\omega^{2}L_{0}} = {{Re}\left\{ \frac{1 + {{j\omega}\quad{R_{S}\left( {C_{C} + C_{S}} \right)}}}{\left( {C_{C} + C_{0}} \right) + {{j\omega}\quad{R_{S}\left\lbrack {{C_{C}C_{S}} + {C_{0}\left( {C_{C} + C_{S}} \right)}} \right\rbrack}}} \right\}}} \\ {= {\frac{\left( {C_{C} + C_{0}} \right) + {\omega^{2}{{R_{S}^{2}\left( {C_{C} + C_{S}} \right)}\left\lbrack {{C_{C}C_{S}} + {C_{0}\left( {C_{C} + C_{S}} \right)}} \right\rbrack}}}{\left( {C_{C} + C_{S}} \right)^{2} + {\omega^{2}\left\lbrack {{C_{C}C_{S}} + {C_{0}\left( {C_{C} + C_{S}} \right)}} \right\rbrack}^{2}}.}} \end{matrix}$

R_(S) is neglected since it is large, so ${{\omega^{2}L_{0}} \approx \frac{\left( {C_{C} + C_{S}} \right)}{{C_{C}C_{S}} + {C_{0}\left( {C_{C} + C_{S}} \right)}}} = {\frac{1}{C_{0}}{\frac{1}{\left\lbrack {1 + \frac{C_{C}C_{S}}{C_{0}\left( {C_{C} + C_{S}} \right)}} \right\rbrack}.}}$ Therefore, $\begin{matrix} {\omega_{0}^{\prime^{2}} = {\frac{1}{L_{0}C_{0}}{\frac{1}{\left\lbrack {1 + \frac{C_{C}C_{S}}{C_{0}\left( {C_{C} + C_{S}} \right)}} \right\rbrack}.}}} & (46) \end{matrix}$

Solving for ω′₀ in equation (46) results in $\begin{matrix} {\omega_{0}^{\prime} = {\omega_{0}{\frac{1}{\sqrt{1 + \frac{C_{C}C_{S}}{C_{0}\left( {C_{C} + C_{S}} \right)}}}.}}} & (47) \end{matrix}$

The Taylor expansion of equation (47) gives $\begin{matrix} {\omega_{0}^{\prime} = {{\omega_{0}\left\lbrack {1 - \frac{C_{C}C_{S}}{2{C_{0}\left( {C_{C} + C_{S}} \right)}}} \right\rbrack}.}} & (48) \end{matrix}$

The sensitivity S_(f) for a dielectric is defined as $\begin{matrix} {{S_{f} = {\frac{g_{S}}{2\pi}{\frac{\mathbb{d}\omega_{0}^{\prime}}{\mathbb{d}C_{S}}}}},} & (49) \end{matrix}$ where ${g_{S} = \frac{A_{eff}}{\xi_{S}}},$ A_(eff) is the effective tip area, and ξ_(S) is the decay length of the evanescent wave, which is approximately 100 μm. Therefore, the sensitivity S_(f) for a dielectric is found by taking the derivative of ω′₀ with respect to C_(S) in equation (48) $\begin{matrix} {S_{f} = {\frac{g_{S}\omega_{0}}{4\pi}{\frac{C_{C}^{2}}{{C_{0}\left( {C_{C} + C_{S}} \right)}^{2}}.}}} & (50) \end{matrix}$

The ability of the probe to differentiate between regions of different permittivity Δ∈/∈ is defined as $\begin{matrix} {\frac{\Delta\quad ɛ}{ɛ} = {{\left( \frac{V_{n{({rms})}}}{V_{i\quad n}} \right)/S_{f}}S_{r}{ɛ.}}} & (51) \end{matrix}$

The experimental verification of the sensitivity for superconductors is performed on a YBa₂Cu₃O_(7-δ) coated SrTiO₃ bi-crystal of 6° orientation mismatch. Resonant frequency shift measurements are taken, resulting in complex permittivity values for two separate locations below T_(c) at 79.4 K. The measurements are taken in the boundary at points C and D shown in FIG. 8. The sensitivities given by equations (14), (24), and (25) are listed in Table II. TABLE II SENSITIVITY AND ASSOCIATED PARAMETERS FOR SUPERCONDUCTORS ε′/ε₀ (10⁸) S_(r) S_(f) Δσ/σ Position C −8.94 9.03 × 10⁻⁶ 1.13 × 10⁻⁶ 1.0 × 10⁻² Position D −8.87 1.04 × 10⁻⁵ 1.13 × 10⁻⁶ 8.6 × 10⁻³

The sensitivity parameters comprise C_(C)=1.36×10⁻¹⁵ F, C₀=8.91×10⁻¹² F, L₀=2.03×10⁻⁸ H, R_(S)=1×10⁻⁶ Ω, σ=3.3×10⁸ S/m, and g_(s)=1.02×10⁻³. The experimental results show that Δσ/σ≅7.8×10⁻³.

The experimental verification of the sensitivity for conductors is also performed on the YBa₂Cu₃O_(7-δ) coated SrTiO₃ bi-crystal of 6° orientation mismatch. The measurements are taken at the same locations for the superconductor sensitivity, in the boundary at points C and D (FIG. 14) at a temperature of 300 K. The sensitivities given by equations (14), (40), and (41) are listed in Table III. The sensitivity parameters consist of C_(C)=1.36×10⁻¹⁵ F, C₀=8.91×10⁻¹² F, L₀=2.03×10⁻⁸ H, R_(S)=7.76×10⁻⁴ Ω[8], σ=1.28×10³ S/m, and g_(c)=1.54×10⁻⁴. The experimental results have shown that Δσ/σ≅2.4×10⁻². TABLE III SENSITIVITY AND ASSOCIATED PARAMETERS FOR CONDUCTORS ε″/ε₀ S_(r) S_(f) Δσ/σ Position C 6.3 6.83 × 10⁻⁶ 5.9 × 10⁻² 8.36 × 10⁻² Position D 6.15 5.95 × 10⁻⁶ 5.9 × 10⁻² 9.91 × 10⁻²

The experimental verification of the sensitivity for dielectrics is performed on single crystal SrTiO₃ utilizing the ferroelectric dependence on temperature property of the material, i.e., ∈_(r)=f(T). The probe tip is set to a 1 μm distance above the sample and tuned to a resonant frequency of 1.114787 GHz at a temperature of 300 K and is illustrated in FIG. 15. The temperature is raised in 0.2 K increments until the resonance shifted in frequency to 1.114792 GHz at 302 K due to the change in dielectric constant and is shown in FIG. 16. The change in dielectric constant is determined using the Curie-Weiss law and results in Δ∈/∈≅6.23×10⁻³. The sensitivity parameters consist of ∈′/∈₀=320.8, C_(C)=1.36×10⁻¹⁵ F, C₀=8.91×10⁻¹² F, C_(S)=4.37×10⁻¹⁵ F, and g_(s)=1.54×10⁻⁶. The lowest theoretically estimated change in permittivity that can be detected by the sensor was Δ∈/∈=5.75×10⁻⁴.

It is noted that terms like “preferably,” “commonly,” and “typically” are not utilized herein to limit the scope of the claimed invention or to imply that certain features are critical, essential, or even important to the structure or function of the claimed invention. Rather, these terms are merely intended to highlight alternative or additional features that may or may not be utilized in a particular embodiment of the present invention.

Having described the invention in detail and by reference to specific embodiments thereof, it will be apparent that modifications and variations are possible without departing from the scope of the invention defined in the appended claims. Accordingly, all variations of the present invention that would readily occur to one of ordinary skill in the art are contemplated to be within the scope of the present invention. Thus, the present invention is not to be limited to the examples and embodiments set forth herein. Rather, the claims alone shall set for the metes and bounds of the present invention. 

1. An evanescent microwave microscopy probe, comprising: a center conductor having a first end and a second end, wherein the center conductor comprises a waveguide for microwave radiation; a probe tip affixed to the first end of the center conductor, wherein the tip is capable of acquiring a near-field microwave signal from a sample; an outer shield surrounding the center conductor, wherein the center conductor and outer shield are in a generally coaxial relationship, wherein the outer shield has a first end and a second end corresponding to the first and second ends of the center conductor, and wherein the center conductor and outer shield are not in direct contact and thereby form a gap; an insulating material occupying at least a portion of the gap between the center conductor and the outer shield; an aperture located near the tip, wherein the aperture comprises a plate having an inside face and an outside face, wherein the aperture is oriented generally perpendicular to the center conductor, and wherein the aperture comprises a hole that allows the tip to be in microwave communication with a sample; and a tuning network in electronic communication with the second end of the center conductor and with the outer shield, wherein the tuning network comprises a pair of capacitors in a parallel electronic relationship.
 2. The probe of claim 1, wherein the insulating material is selected from paraffin, magnesium oxide, titanium oxide, boron nitride, alumina, an organic polymer, or any combination of two or more thereof.
 3. The probe of claim 1, wherein the aperture further comprises copper or a copper alloy.
 4. The probe of claim 3, wherein the aperture is affixed to the outer shield with solder.
 5. The probe of claim 1, wherein the tuning network further comprises sapphire capacitors.
 6. The probe of claim 1, wherein the inside face of the aperture includes a chamfered surface about the circumference defining the hole.
 7. The probe of claim 6, wherein a ceramic coating is disposed on the chamfered surface.
 8. A process for making a microwave probe comprising: providing a center conductor having a first end and a second end, wherein the center conductor comprises a waveguide for microwave radiation; affixing a probe tip to the first end of the center conductor, wherein the tip is capable of acquiring a near-field microwave signal from a sample; surrounding the center conductor with an outer shield, wherein the center conductor and outer shield are in a generally coaxial relationship, wherein the outer shield has a first end and a second end corresponding to the first and second ends of the center conductor, and wherein the center conductor and outer shield are not in direct contact and thereby form a gap; occupying at least a portion of the gap between the center conductor and the outer shield with an insulating material; providing an aperture located near the tip, wherein the aperture comprises a plate having an inside face and an outside face, wherein the aperture is oriented generally perpendicular to the center conductor, and wherein the aperture comprises a hole that allows the tip to be in microwave communication with a sample; and providing a tuning network in electronic communication with the second end of the center conductor and with the outer shield, wherein the tuning network comprises a pair of capacitors in a parallel electronic relationship.
 9. The probe of claim 8, wherein the insulating material is selected from paraffin, magnesium oxide, titanium oxide, boron nitride, alumina, an organic polymer, or any combination of two or more thereof.
 10. The probe of claim 8, wherein the aperture further comprises copper or a copper alloy.
 11. The probe of claim 10, wherein the aperture is affixed to the outer shield with solder.
 12. The probe of claim 8, wherein the tuning network further comprises sapphire capacitors.
 13. The probe of claim 8, wherein the inside face of the aperture includes a chamfered surface about the circumference defining the hole.
 14. The probe of claim 13, wherein a ceramic coating is disposed on the chamfered surface.
 15. A method for detecting a sample using an electromagnetic microwave field comprising: providing the probe of claim 1; obtaining a resonant frequency reference reading from the probe, wherein the probe is substantially decoupled from a sample; placing the probe of claim 1 in electromagnetic microwave communication with the sample; obtaining an resonant frequency reading from the probe; calculating a resonant frequency change relative to the reference reading; and relating the resonant frequency change to one or more properties of the sample.
 16. A method for imaging a sample using an electromagnetic microwave field comprising: providing the probe of claim 1; providing an X-Y sample stage, and a sample disposed thereon; obtaining a resonant frequency reference reading from the probe, wherein the probe is substantially decoupled from the sample; placing the probe of claim 1 in electromagnetic microwave communication with the sample at a first position; obtaining a resonant frequency reading from the probe at the first position; moving the probe to a next position; obtaining a resonant frequency reading from the probe at the next position; repeating the preceding two steps as needed; calculating a resonant frequency change at each position relative to the reference reading; and plotting an image of the sample as a function of position and frequency change or a property related to frequency change. 